1. Field of the Invention
The present invention relates to a correcting circuit for a mixer circuit, to a receiver using such a correcting circuit, and to a frequency spectrum-inverting circuit using such a correcting circuit. More particularly, the invention relates to a correcting circuit which is for use with a mixing circuit and acts to reduce carrier leakage, to a receiver and a frequency spectrum-inverting circuit using such a correcting circuit.
2. Background of the Invention
One known superheterodyne receiver is a double superheterodyne, direct-conversion type receiver as shown in FIG. 1. This receiver performs its first and second frequency conversions, utilizing orthogonal conversion, to improve the image rejection characteristics. This receiver operates as follows.
The receiver shown in FIG. 1 constitutes the receiver portion of a cordless telephone set, and receives an FM signal. The FM signal, Sr, to be received is applied to its input terminal 1 and fed to first mixer circuits 11 and 21, for I-axis and Q-axis, respectively, for orthogonal conversion via an RF amplifier 2.
A first local oscillator circuit 14 produces a first local oscillator signal S.sub.14 having the same frequency as the carrier frequency of the received signal Sr. This signal S.sub.14 is applied to the mixer circuit 11 and also to a phase-shift circuit 24, where the signal is shifted in phase by .pi./2, resulting in a phase-shifted signal S.sub.24. This signal S.sub.24 is supplied as the first local oscillator signal to the mixer circuit 21.
For simplicity, it is assumed that the received signal Sr has a signal component Sa within its lower sideband and a signal component Sb within its upper sideband, as shown in FIG. 2A, where .omega..sub.0 is the carrier frequency (angular frequency) of the received signal Sr, .omega..sub.a, is the angular frequency (.omega..sub.a &lt;.omega..sub.0) of the signal component Sa, Ea is the amplitude of the signal component Sa, .omega..sub.b is the angular frequency (.omega..sub.a &gt;.omega..sub.0) of the signal component Sb, E.sub.b is the amplitude of the signal component S.sub.b, .DELTA..omega..sub.a =.omega..sub.0 -.omega..sub.a, and .DELTA..omega..sub.b =.omega..sub.b -.omega..sub.0. We can have the relations EQU Sr=Sa+Sb EQU Sa=Ea.multidot.sin .omega..sub.a t EQU Sb=Eb.multidot.sin .omega..sub.b t
Assuming that E.sub.1 is the amplitude of the first local oscillator signals S.sub.14 and S.sub.24, we have the relationships EQU S.sub.14 =E.sub.1 .multidot.sin .omega..sub.0 t EQU S.sub.24 =E.sub.1 .multidot.cos .omega..sub.0 t
Let S.sub.11 and S.sub.12 be the output signals from the mixer circuits 11 and 21, respectively. These output signals can be expressed in the manner described below. ##EQU1## Of the above formulas, the signal components of the angular frequencies .DELTA..omega..sub.a and .DELTA..omega..sub.b are desired intermediate frequencies and so these signals S.sub.11 and S.sub.21 are supplied to low-pass filters 12 and 22, respectively. The signal components of the angular frequencies .DELTA..omega..sub.a and .DELTA..omega..sub.b are taken out as first intermediate-frequency signals S.sub.12 and S.sub.22, respectively. Thus, we have EQU S.sub.12 =.alpha.a.multidot.cos .DELTA..omega..sub.a t+.alpha.b.multidot.cos .DELTA..omega..sub.b t EQU S.sub.22 =-.alpha.a.multidot.sin .DELTA..omega..sub.a t+.alpha.b.multidot.sin .DELTA..omega..sub.b t
In this case, as can be seen from FIG. 2A, the signals S.sub.12 and S.sub.22, are signals in the baseband.
These signals S.sub.12 and S.sub.22 are supplied to second mixer circuits 13 and 23, respectively, for I-axis and Q-axis, respectively, for orthogonal conversion. A comparatively low-frequency, second local oscillator signal S.sub.15 is taken from a second local oscillator circuit 15. This signal S.sub.15 is fed to the mixer circuit 13 and also to a phase-shift circuit 25, where the signal is shifted in phase by .pi./2. The phase-shifted signal is supplied as the second local oscillator signal to the mixer circuit 23. Accordingly, we have EQU S.sub.15 =E.sub.2 .multidot.sin .omega..sub.s t EQU S.sub.25 =E.sub.2 .multidot.cos .omega..sub.s t
where E.sub.2, is the amplitude of the second local oscillator signals S.sub.15 and S.sub.25 and .omega..sub.s =2.pi.f.sub.s. For example, f.sub.s is 55 kHz. Let S.sub.13 and S.sub.14 be the output signals from the mixer circuits 13 and 23, respectively. Thus, ##EQU2## By modifying these signals S.sub.13 and S.sub.23 in such a way that any frequency difference does not assume a negative value, we have ##EQU3## These signals S.sub.13 and S.sub.23 are supplied to an adder circuit 3, where they are summed up. The adder circuit 3 produces a sum signal given by ##EQU4## This sum signal S.sub.3 is illustrated in FIG. 2B. This signal S.sub.3 is obtained by converting the original signal Sr into a signal of the carrier frequency, or the angular frequency .omega..sub.s. That is, the signal S.sub.3 is a second intermediate-frequency signal of intermediate frequency f.sub.s.
This second intermediate-frequency signal S.sub.3 is supplied via an IF bandpass filter 4 and via a limiter amplifier 5 to an FM demodulator circuit 6, where the original signal is demodulated. This audio signal appears at an output terminal 7.
If the adder circuit 3 does not produce the sum of the signals S.sub.13 and S.sub.23 but performs a subtraction, then ##EQU5## This signal Simg is a signal having a distribution that is an inversion of the frequency spectrum of the above-described original second intermediate-frequency signal S.sub.3 in the frequency band occupied by this signal S.sub.3, i.e., the signal Simg is an image interference signal.
In an ordinary FM receiver, this intermediate frequency is set to 10.7 MHz. Therefore, the intermediate-frequency filter is inevitably made of a ceramic filter. This makes it impossible to fabricate the circuit as an IC. In the above-described receiver, however, the first intermediate-frequency (IF) signals S.sub.12 and S.sub.22 are in the baseband. Consequently, the second IF frequency f.sub.s can be set to a low value, e.g., 55 kHz. Hence, each of the filters 12, 22, and 4 can consist of an active filter made up of resistors, capacitors, and an amplifier. Thus, the portions from the terminal 1 to the terminal 7, including the filters 12, 22, and 4, can be integrated into one monolithic IC chip.
The receiver of the construction described above cannot be put into practical use because it has two great problems. The first one arises because the second mixer circuits 13 and 23 do not operate ideally. In particular, some of the second local oscillator signals S.sub.15 and S.sub.25 leak to the adder circuit 3 through the mixer circuits 13 and 23, thus causing trouble.
More specifically, each of the mixer circuits 13 and 23 is generally made of a double balanced switching circuit, or a balanced modulator circuit, as shown in FIG. 3. The first IF signal S.sub.12 or S.sub.22 is supplied across the bases of lower transistors Q.sub.2 and Q.sub.3. The rectangular second local oscillator signal S.sub.15 or S.sub.25 is supplied across the bases of upper transistors Q.sub.4 and Q.sub.5 and across the bases of upper transistors Q.sub.6 and Q.sub.7. The signal S.sub.12 produced from the collectors of the transistors Q.sub.2 and Q.sub.3 is chopped by the signal S.sub.15. The second IF signal S.sub.13 or S.sub.23 is taken from the collectors of the transistors Q.sub.4 -Q.sub.7.
In this case, if the characteristics of the transistors Q.sub.1 -Q.sub.7 and their DC biases are completely balanced against each other, then only the desired second IF signal S.sub.13 is produced from the transistors Q.sub.4 -Q.sub.7. In practice, however, the characteristics of the transistors Q.sub.1 -Q.sub.7 and their DC biases are not matched, in which case the second IF signal S.sub.13 produced from the transistors Q.sub.4 -Q.sub.7 contains a carrier component, or a component of the second local oscillator signal S.sub.15. That is, carrier leakage occurs.
Accordingly, it is common practice to adjust the bias voltage V.sub.B3 applied to the base of the transistor Q.sub.3 so as to minimize the carrier leakage, based on the voltage V.sub.B2 applied to the base of the transistor Q.sub.2, as shown in FIG. 3.
In this method, each individual receiver is required to be adjusted. This deteriorates the productivity of receivers. If the ambient temperature around the receiver varies, the optimally adjusted point changes and so temperature compensation is needed. This compensation is not easy to perform.
One effective method of maintaining the accuracy of the balance between the characteristics of the transistors and between the DC biases is to fabricate components as an IC. However, if the mixer circuits 13 and 23 shown in FIG. 3 is fabricated as an IC, carrier can be suppressed to about 40-50 dB at best. If the carrier leakage is suppressed further, external adjustment is yet needed.
If the carrier signal leaks from the mixer circuits 13 and 23 in this way, the carrier frequency is f.sub.2 and equal to the second intermediate frequency f.sub.2 in the case of the receiver shown in FIG. 1. Therefore, it is very difficult to separate the second IF signal S.sub.3 from the leaking carrier component. As a result, if automatic gain control (AGC) is provided, the level of the received electric field is displayed, and tuning is indicated, then malfunctions will occur. Furthermore, the leaking carrier component beats with the second IF signal S.sub.3, thus deteriorating the reception characteristics.
The second problem arises from the fact that the first IF signals S.sub.12 and S.sub.22 are in the baseband. That is, if the received signal Sr has a large level, then it cannot be treated. More specifically, in the receiver shown in FIG. 1, in the signal intervals between the first mixer circuits 11, 21 and the second mixer circuits 13, 23, respectively, the following relation holds: EQU first intermediate frequency.ltoreq.frequencies of signal components
Therefore, even where the received signal Sr is an FM signal, if the first IF signals S.sub.12 and S.sub.22 are clipped, information is lost during this interval. It follows that the signals S.sub.12 and S.sub.22 are sampled at a frequency lower than the frequencies of these signals. As a result, beating and distortions take place.